Dynamic igbt gate drive to reduce switching loss

ABSTRACT

A vehicle includes an electric machine, an IGBT, and a gate driver. The IGBT has a gate, an emitter, and a collector and is configured to flow an electric charge through a phase of the electric machine. The gate driver is configured to flow current onto the gate at a first level, and in response to a time integral of a voltage across the phase exceeding a predetermined level, transition from the first level to a second level less than the first level.

TECHNICAL FIELD

This application is generally related to control of a gate current to anIGBT in a hybrid-electric powertrain in which the gate current has atleast two constant current levels.

BACKGROUND

Electrified vehicles including hybrid-electric vehicles (HEVs) andbattery electric vehicles (BEVs) rely on a traction battery to providepower to a traction motor for propulsion and a power invertertherebetween to convert direct current (DC) power to alternating current(AC) power. The typical AC traction motor is a 3-phase motor that may bepowered by 3 sinusoidal signals each driven with 120 degrees phaseseparation. The traction battery is configured to operate in aparticular voltage range. The terminal voltage of a typical tractionbattery is over 100 Volts DC, and the traction battery is alternativelyreferred to as a high-voltage battery. However, improved performance ofelectric machines may be achieved by operating in a different voltagerange, typically at higher voltages than the traction battery.

Many electrified vehicles include a DC-DC converter also referred to asa variable voltage converter (VVC) to convert the voltage of thetraction battery to an operational voltage level of the electricmachine. The electric machine that may include a traction motor mayrequire a high voltage and high current. Due to the voltage, current andswitching requirements, an Insulated Gate Bipolar junction Transistor(IGBT) is typically used to generate the signals in the power inverterand the VVC.

SUMMARY

A vehicle includes an electric machine, an IGBT, and a gate driver. TheIGBT has a gate, an emitter, and a collector and is configured to flowan electric charge through a phase of the electric machine. The gatedriver is configured to flow current onto the gate at a first level, andin response to a time integral of a voltage across the phase exceeding apredetermined level, transition from the first level to a second levelless than the first level.

A method of controlling an IGBT of a vehicle powertrain includes flowinga current at a first level onto a gate of an IGBT, and in response to atime integral of a voltage across the IGBT exceeding a predeterminedthreshold, transitioning the current from the first level to a secondlevel less than the first level.

A vehicle powertrain includes an IGBT and a gate driver. The IGBT has agate, an emitter and a collector. The gate driver is configured to flowcurrent onto the gate at a first level, and in response to a timeintegral of resulting collector to emitter voltage exceeding apredetermined level, transition from the first level to a second levelless than the first level.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram of a hybrid vehicle illustrating typical drivetrainand energy storage components with a power inverter therebetween.

FIG. 2 is a schematic diagram of a vehicular variable voltage converter.

FIG. 3 is a schematic diagram of a vehicular electric motor inverter.

FIG. 4 is a graphical representation of a gate current with respect totime.

FIG. 5 is a flow diagram of a method to drive a gate of an IGBT.

FIG. 6 is a schematic diagram of an IGBT gate drive circuit.

DETAILED DESCRIPTION

Embodiments of the present disclosure are described herein. It is to beunderstood, however, that the disclosed embodiments are merely examplesand other embodiments can take various and alternative forms. Thefigures are not necessarily to scale; some features could be exaggeratedor minimized to show details of particular components. Therefore,specific structural and functional details disclosed herein are not tobe interpreted as limiting, but merely as a representative basis forteaching one skilled in the art to variously employ the presentinvention. As those of ordinary skill in the art will understand,various features illustrated and described with reference to any one ofthe figures can be combined with features illustrated in one or moreother figures to produce embodiments that are not explicitly illustratedor described. The combinations of features illustrated providerepresentative embodiments for typical applications. Variouscombinations and modifications of the features consistent with theteachings of this disclosure, however, could be desired for particularapplications or implementations.

Semiconductor devices such as metal-oxide-semiconductor field-effecttransistor (MOSFET) or Insulated Gate Bipolar junction Transistors(IGBTs) and flyback or freewheeling diodes are widely used in a varietyof power systems including but not limited to consumer, medical, andindustrial applications, such as electric motor drives and powerinverters. Here, gate control of an IGBT is illustrated; however, theconcepts and structures are also applicable to MOSFETs Operation of anIGBT is controlled by a gate voltage supplied by a gate driver.Conventional gate drivers are typically based on a voltage, greater thana threshold voltage, applied to an IGBT gate with a current limitingresistor, which consists of a switchable voltage source and gateresistor. A low gate resistance would lead to a fast switching speed andlow switching loss, but also cause higher stresses on the semiconductordevices, e.g. over-voltage stress. Therefore, the gate resistance isselected to seek a compromise between switching loss, switching delay,and stresses.

Some disadvantages associated with conventional gate drivers for IGBTturn-on include limited control of switching delay time, current slopeand voltage slope such that optimization switching losses is limited.Another disadvantage is that a gate resistance is typically selectedbased on worst case operating condition thus introducing excessiveswitching losses under normal operating conditions. For example, at ahigh dc bus voltage, a gate resistance is selected based on a change incurrent with respect to time (di/dt) in order to avoid excessive diodevoltage overshoot during diode fly-back of the load. However, at low dcbus voltage the use of the gate resistance selected to protect for highbus voltages introduces excessive switching losses as a switching speedis then reduced by the gate resistance even though diode over-voltage isbelow a threshold.

A smart gate driving strategy is critical to achieve optimal switchingperformance for the whole switching trajectory and over all theoperating ranges. Here, a proposed stepwise current source gate drivingstrategy with feedback of operating conditions (e.g., voltage, loadcurrent, temperature, etc.) for IGBT turn-on is presented. In oneembodiment, a two-step gate driving profile is composed of a highcurrent pulse (Ig1) driven for a first predetermined time followed by alow current pulse (Ig2) driven for a second predetermined time. The highcurrent pulse is selected to reduce the turn-on delay time, as well asincrease switching speed and reduce switching loss. The low currentpulse slows down the switching speed to avoid the excessive voltageovershoot across the associated freewheeling diode. The timing for eachpulse stage is adaptive to IGBT operating conditions, e.g., switchedvoltage (Vce), to realize the optimal switching performance over thewhole operating ranges. The gate driver produces the longest Ig1possible based on the operating conditions, in order to achieve aminimum switching loss while keeping the diode voltage overshoot withina safety limit. At low voltage cases, Ig1 could go all the way throughthe turn on transient, allowing a fast completion of switching transientto avoid excessive turn-on loss.

FIG. 1 depicts an electrified vehicle 112 that may be referred to as aplug-in hybrid-electric vehicle (PHEV). A plug-in hybrid-electricvehicle 112 may comprise one or more electric machines 114 mechanicallycoupled to a hybrid transmission 116. The electric machines 114 may becapable of operating as a motor or a generator. In addition, the hybridtransmission 116 is mechanically coupled to an engine 118. The hybridtransmission 116 is also mechanically coupled to a drive shaft 120 thatis mechanically coupled to the wheels 122. The electric machines 114 canprovide propulsion and deceleration capability when the engine 118 isturned on or off. The electric machines 114 may also act as generatorsand can provide fuel economy benefits by recovering energy that wouldnormally be lost as heat in a friction braking system. The electricmachines 114 may also reduce vehicle emissions by allowing the engine118 to operate at more efficient speeds and allowing the hybrid-electricvehicle 112 to be operated in electric mode with the engine 118 offunder certain conditions. An electrified vehicle 112 may also be abattery electric vehicle (BEV). In a BEV configuration, the engine 118may not be present. In other configurations, the electrified vehicle 112may be a full hybrid-electric vehicle (FHEV) without plug-in capability.

A traction battery or battery pack 124 stores energy that can be used bythe electric machines 114. The vehicle battery pack 124 may provide ahigh voltage direct current (DC) output. The traction battery 124 may beelectrically coupled to one or more power electronics modules 126. Oneor more contactors 142 may isolate the traction battery 124 from othercomponents when opened and connect the traction battery 124 to othercomponents when closed. The power electronics module 126 is alsoelectrically coupled to the electric machines 114 and provides theability to bi-directionally transfer energy between the traction battery124 and the electric machines 114. For example, a traction battery 124may provide a DC voltage while the electric machines 114 may operatewith a three-phase alternating current (AC) to function. The powerelectronics module 126 may convert the DC voltage to a three-phase ACcurrent to operate the electric machines 114. In a regenerative mode,the power electronics module 126 may convert the three-phase AC currentfrom the electric machines 114 acting as generators to the DC voltagecompatible with the traction battery 124.

The vehicle 112 may include a variable-voltage converter (VVC) 152electrically coupled between the traction battery 124 and the powerelectronics module 126. The VVC 152 may be a DC/DC boost converterconfigured to increase or boost the voltage provided by the tractionbattery 124. By increasing the voltage, current requirements may bedecreased leading to a reduction in wiring size for the powerelectronics module 126 and the electric machines 114. Further, theelectric machines 114 may be operated with better efficiency and lowerlosses.

In addition to providing energy for propulsion, the traction battery 124may provide energy for other vehicle electrical systems. The vehicle 112may include a DC/DC converter module 128 that converts the high voltageDC output of the traction battery 124 to a low voltage DC supply that iscompatible with low-voltage vehicle loads. An output of the DC/DCconverter module 128 may be electrically coupled to an auxiliary battery130 (e.g., 12V battery) for charging the auxiliary battery 130. Thelow-voltage systems may be electrically coupled to the auxiliary battery130. One or more electrical loads 146 may be coupled to the high-voltagebus. The electrical loads 146 may have an associated controller thatoperates and controls the electrical loads 146 when appropriate.Examples of electrical loads 146 may be a fan, an electric heatingelement and/or an air-conditioning compressor.

The electrified vehicle 112 may be configured to recharge the tractionbattery 124 from an external power source 136. The external power source136 may be a connection to an electrical outlet. The external powersource 136 may be electrically coupled to a charger or electric vehiclesupply equipment (EVSE) 138. The external power source 136 may be anelectrical power distribution network or grid as provided by an electricutility company. The EVSE 138 may provide circuitry and controls toregulate and manage the transfer of energy between the power source 136and the vehicle 112. The external power source 136 may provide DC or ACelectric power to the EVSE 138. The EVSE 138 may have a charge connector140 for plugging into a charge port 134 of the vehicle 112. The chargeport 134 may be any type of port configured to transfer power from theEVSE 138 to the vehicle 112. The charge port 134 may be electricallycoupled to a charger or on-board power conversion module 132. The powerconversion module 132 may condition the power supplied from the EVSE 138to provide the proper voltage and current levels to the traction battery124. The power conversion module 132 may interface with the EVSE 138 tocoordinate the delivery of power to the vehicle 112. The EVSE connector140 may have pins that mate with corresponding recesses of the chargeport 134. Alternatively, various components described as beingelectrically coupled or connected may transfer power using a wirelessinductive coupling.

One or more wheel brakes 144 may be provided for decelerating thevehicle 112 and preventing motion of the vehicle 112. The wheel brakes144 may be hydraulically actuated, electrically actuated, or somecombination thereof. The wheel brakes 144 may be a part of a brakesystem 150. The brake system 150 may include other components to operatethe wheel brakes 144. For simplicity, the figure depicts a singleconnection between the brake system 150 and one of the wheel brakes 144.A connection between the brake system 150 and the other wheel brakes 144is implied. The brake system 150 may include a controller to monitor andcoordinate the brake system 150. The brake system 150 may monitor thebrake components and control the wheel brakes 144 for vehicledeceleration. The brake system 150 may respond to driver commands andmay also operate autonomously to implement features such as stabilitycontrol. The controller of the brake system 150 may implement a methodof applying a requested brake force when requested by another controlleror sub-function.

Electronic modules in the vehicle 112 may communicate via one or morevehicle networks. The vehicle network may include a plurality ofchannels for communication. One channel of the vehicle network may be aserial bus such as a Controller Area Network (CAN). One of the channelsof the vehicle network may include an Ethernet network defined byInstitute of Electrical and Electronics Engineers (IEEE) 802 family ofstandards. Additional channels of the vehicle network may includediscrete connections between modules and may include power signals fromthe auxiliary battery 130. Different signals may be transferred overdifferent channels of the vehicle network. For example, video signalsmay be transferred over a high-speed channel (e.g., Ethernet) whilecontrol signals may be transferred over CAN or discrete signals. Thevehicle network may include any hardware and software components thataid in transferring signals and data between modules. The vehiclenetwork is not shown in FIG. 1 but it may be implied that the vehiclenetwork may connect to any electronic module that is present in thevehicle 112. A vehicle system controller (VSC) 148 may be present tocoordinate the operation of the various components.

FIG. 2 depicts a diagram of a VVC 152 that is configured as a boostconverter. The VVC 152 may include input terminals that may be coupledto terminals of the traction battery 124 through the contactors 142. TheVVC 152 may include output terminals coupled to terminals of the powerelectronics module 126. The VVC 152 may be operated to cause a voltageat the output terminals to be greater than a voltage at the inputterminals. The vehicle 112 may include a VVC controller 200 thatmonitors and controls electrical parameters (e.g., voltage and current)at various locations within the VVC 152. In some configurations, the VVCcontroller 200 may be included as part of the VVC 152. The VVCcontroller 200 may determine an output voltage reference, V_(dc)*. TheVVC controller 200 may determine, based on the electrical parameters andthe voltage reference, V_(dc)*, a control signal sufficient to cause theVVC 152 to achieve the desired output voltage. In some configurations,the control signal may be implemented as a pulse-width modulated (PWM)signal in which a duty cycle of the PWM signal is varied. The controlsignal may be operated at a predetermined switching frequency. The VVCcontroller 200 may command the VVC 152 to provide the desired outputvoltage using the control signal. The particular control signal at whichthe VVC 152 is operated may be directly related to the amount of voltageboost to be provided by the VVC 152.

The output voltage of the VVC 152 may be controlled to achieve a desiredreference voltage. In some configurations, the VVC 152 may be a boostconverter. In a boost converter configuration in which the VVCcontroller 200 controls the duty cycle, the ideal relationship betweenthe input voltage V_(in) and the output voltage V_(out) and the dutycycle D may be illustrated using the following equation:

$\begin{matrix}{V_{out} = \frac{V_{in}}{\left( {1 - D} \right)}} & \left. 1 \right)\end{matrix}$

The desired duty cycle, D, may be determined by measuring the inputvoltage (e.g., traction battery voltage) and setting the output voltageto the reference voltage. The VVC 152 may be a buck converter thatreduces the voltage from input to output. In a buck configuration, adifferent expression relating the input and output voltage to the dutycycle may be derived. In some configurations, the VVC 152 may be abuck-boost converter that may increase or decrease the input voltage.The control strategy described herein is not limited to a particularvariable voltage converter topology.

With reference to FIG. 2, the VVC 152 may boost or “step up” the voltagepotential of the electrical power provided by the traction battery 124.The traction battery 124 may provide high voltage (HV) DC power. In someconfigurations, the traction battery 124 may provide a voltage between150 and 400 Volts. The contactor 142 may be electrically coupled inseries between the traction battery 124 and the VVC 152. When thecontactor 142 is closed, the HV DC power may be transferred from thetraction battery 124 to the VVC 152. An input capacitor 202 may beelectrically coupled in parallel to the traction battery 124. The inputcapacitor 202 may stabilize the bus voltage and reduce any voltage andcurrent ripple. The VVC 152 may receive the HV DC power and boost or“step up” the voltage potential of the input voltage according to theduty cycle.

An output capacitor 204 may be electrically coupled between the outputterminals of the VVC 152. The output capacitor 204 may stabilize the busvoltage and reduce voltage and current ripple at the output of the VVC152.

Further with reference to FIG. 2, the VVC 152 may include a firstswitching device 206 and a second switching device 208 for boosting aninput voltage to provide the boosted output voltage. The switchingdevices 206, 208 may be configured to selectively flow a current to anelectrical load (e.g., power electronics module 126 and electricmachines 114). Each switching device 206, 208 may be individuallycontrolled by a gate drive circuit (not shown) of the VVC controller 200and may include any type of controllable switch (e.g., an insulated gatebipolar transistor (IGBT) or field-effect transistor (FET)). The gatedrive circuit may provide electrical signals to each of the switchingdevices 206, 208 that are based on the control signal (e.g., duty cycleof PWM control signal). A diode may be coupled across each of theswitching devices 206, 208. The switching devices 206, 208 may each havean associated switching loss. The switching losses are those powerlosses that occur during state changes of the switching device (e.g.,on/off and off/on transitions). The switching losses may be quantifiedby the current flowing through and the voltage across the switchingdevice 206, 208 during the transition. The switching devices may alsohave associated conduction losses that occur when the device is switchedon.

The vehicle system may include sensors for measuring electricalparameters of the VVC 152. A first voltage sensor 210 may be configuredto measure the input voltage, (e.g., voltage of the battery 124), andprovide a corresponding input signal (V_(bat)) to the VVC controller200. In one or more embodiments, the first voltage sensor 210 maymeasure the voltage across the input capacitor 202, which corresponds tothe battery voltage. A second voltage sensor 212 may measure the outputvoltage of the VVC 152 and provide a corresponding input signal (V_(dc))to the VVC controller 200. In one or more embodiments, the secondvoltage sensor 212 may measure the voltage across the output capacitor204, which corresponds to the DC bus voltage. The first voltage sensor210 and the second voltage sensor 212 may include circuitry to scale thevoltages to a level appropriate for the VVC controller 200. The VVCcontroller 200 may include circuitry to filter and digitize the signalsfrom the first voltage sensor 210 and the second voltage sensor 212.

An input inductor 214 may be electrically coupled in series between thetraction battery 124 and the switching devices 206, 208. The inputinductor 214 may alternate between storing and releasing energy in theVVC 152 to enable the providing of the variable voltages and currents asVVC 152 output, and the achieving of the desired voltage boost. Acurrent sensor 216 may measure the input current through the inputinductor 214 and provide a corresponding current signal (I_(L)) to theVVC controller 200. The input current through the input inductor 214 maybe a result of the voltage difference between the input and the outputvoltage of the VVC 152, the conducting time of the switching devices206, 208, and the inductance L of the input inductor 214. The VVCcontroller 200 may include circuitry to scale, filter, and digitize thesignal from the current sensor 216.

The VVC controller 200 may be programmed to control the output voltageof the VVC 152. The VVC controller 200 may receive input from the VVC152 and other controllers via the vehicle network, and determine thecontrol signals. The VVC controller 200 may monitor the input signals(V_(bat), V_(dc), I_(L), V_(dc)*) to determine the control signals. Forexample, the VVC controller 200 may provide control signals to the gatedrive circuit that correspond to a duty cycle command. The gate drivecircuit may then control each switching device 206, 208 based on theduty cycle command.

The control signals to the VVC 152 may be configured to drive theswitching devices 206, 208 at a particular switching frequency. Withineach cycle of the switching frequency, the switching devices 206, 208may be operated at the specified duty cycle. The duty cycle defines theamount of time that the switching devices 206, 208 are in an on-stateand an off-state. For example, a duty cycle of 100% may operate theswitching devices 206, 208 in a continuous on-state with no turn off. Aduty cycle of 0% may operate the switching devices 206, 208 in acontinuous off-state with no turn on. A duty cycle of 50% may operatethe switching devices 206, 208 in an on-state for half of the cycle andin an off-state for half of the cycle. The control signals for the twoswitches 206, 208 may be complementary. That is, the control signal sentto one of the switching devices (e.g., 206) may be an inverted versionof the control signal sent to the other switching device (e.g., 208).

The current that is controlled by the switching devices 206, 208 mayinclude a ripple component that has a magnitude that varies with amagnitude of the current, and the duty cycle and switching frequency ofthe switching devices 206, 208. Relative to the input current, the worstcase ripple current magnitude occurs during relatively high inputcurrent conditions. When the duty cycle is fixed, an increase in theinductor current causes an increase in magnitude of the ripple currentas illustrated in FIG. 4. The magnitude of the ripple current is alsorelated to the duty cycle. The highest magnitude ripple current occurswhen the duty cycle equals 50%. The general relationship between theinductor ripple current magnitude and the duty cycle may be as shown inFIG. 5. Based on these facts, it may be beneficial to implement measuresto reduce the ripple current magnitude under high current and mid-rangeduty cycle conditions.

When designing the VVC 152, the switching frequency and the inductancevalue of the inductor 214 may be selected to satisfy a maximum allowableripple current magnitude. The ripple component may be a periodicvariation that appears on a DC signal. The ripple component may bedefined by a ripple component magnitude and a ripple componentfrequency. The ripple component may have harmonics that are in anaudible frequency range that may add to the noise signature of thevehicle. Further, the ripple component may cause difficulties withaccurately controlling devices fed by the source. During switchingtransients, the switching devices 206, 208 may turn off at the maximuminductor current (DC current plus ripple current) which may cause largevoltage spike across the switching devices 206, 208. Because of size andcost constraints, the inductance value may be selected based on theconducted current. In general, as current increases the inductance maydecrease due to saturation.

The switching frequency may be selected to limit a magnitude of theripple current component under worst case scenarios (e.g., highest inputcurrent and/or duty cycle close to 50% conditions). The switchingfrequency of the switching devices 206, 208 may be selected to be afrequency (e.g., 10 kHz) that is greater than a switching frequency ofthe motor/generator inverter (e.g., 5 kHz) that is coupled to an outputof the VVC 152. In some applications, the switching frequency of the VVC152 may be selected to be a predetermined fixed frequency. Thepredetermined fixed frequency is generally selected to satisfy noise andripple current specifications. However, the choice of the predeterminedfixed frequency may not provide best performance over all operatingranges of the VVC 152. The predetermined fixed frequency may providebest results at a particular set of operating conditions, but may be acompromise at other operating conditions.

Increasing the switching frequency may decrease the ripple currentmagnitude and lower voltage stress across the switching devices 206,208, but may lead to higher switching losses. While the switchingfrequency may be selected for worst case ripple conditions, the VVC 152may only operate under the worst case ripple conditions for a smallpercentage of the total operating time. This may lead to unnecessarilyhigh switching losses that may lower fuel economy. In addition, thefixed switching frequency may concentrate the noise spectrum in a verynarrow range. The increased noise density in this narrow range mayresult in noticeable noise, vibration, and harshness (NVH) issues.

The VVC controller 200 may be programmed to vary the switching frequencyof the switching devices 206, 208 based on the duty cycle and the inputcurrent. The variation in switching frequency may improve fuel economyby reducing switching losses and reduce NVH issues while maintainingripple current targets under worst case operating conditions.

During relatively high current conditions, the switching devices 206,208 may experience increased voltage stress. At a maximum operatingcurrent of the VVC 152, it may be desired to select a relatively highswitching frequency that reduces the ripple component magnitude with areasonable level of switching losses. The switching frequency may beselected based on the input current magnitude such that as the inputcurrent magnitude increases, the switching frequency increases. Theswitching frequency may be increased up to a predetermined maximumswitching frequency. The predetermined maximum switching frequency maybe a level that provides a compromise between lower ripple componentmagnitudes and higher switching losses. The switching frequency may bechanged in discrete steps or continuously over the operating currentrange.

The VVC controller 200 may be programmed to reduce the switchingfrequency in response to the current input being less than apredetermined maximum current. The predetermined maximum current may bea maximum operating current of the VVC 152. The change in the switchingfrequency may be based on the magnitude of the current input to theswitching devices 206, 208. When the current is greater than thepredetermined maximum current, the switching frequency may be set to apredetermined maximum switching frequency. As the current decreases, themagnitude of the ripple component decreases. By operating at lowerswitching frequencies as the current decreases, switching losses arereduced. The switching frequency may be varied based on the power inputto the switching devices. As the input power is a function of the inputcurrent and the battery voltage, the input power and input current maybe used in a similar manner.

Since the ripple current is also affected by the duty cycle, theswitching frequency may be varied based on the duty cycle. The dutycycle may be determined based on a ratio of the input voltage to theoutput voltage. As such, the switching frequency may also be variedbased on the ratio between the input voltage and the output voltage.When the duty cycle is near 50%, the predicted ripple current magnitudeis a maximum value and the switching frequency may be set to thepredetermined maximum frequency. The predetermined maximum frequency maybe a maximum switching frequency value that is selected to minimize theripple current magnitude. The switching frequency may be changed indiscrete steps or continuously over the duty cycle range.

The VVC controller 200 may be programmed to reduce the switchingfrequency from the predetermined maximum frequency in response to amagnitude of a difference between the duty cycle and the duty cyclevalue (e.g, 50%) at which the predicted ripple component magnitude is amaximum. When the magnitude of the difference is less than a threshold,the switching frequency may be set to the predetermined frequency. Whenthe magnitude of the difference decreases, the switching frequency maybe increased toward the predetermined maximum frequency to reduce theripple component magnitude. When the magnitude of the difference is lessthan a threshold, the switching frequency may be set to thepredetermined maximum frequency.

The switching frequency may be limited to be between the predeterminedmaximum frequency and a predetermined minimum frequency. Thepredetermined minimum frequency may be a frequency level that is greaterthan a predetermined switching frequency of the power electronic module126 that is coupled to an output of the variable voltage converter 152.The switching frequency may also be based on parasitic inductanceassociated with the gate of the IGBT.

With reference to FIG. 3, a system 300 is provided for controlling apower electronics module (PEM) 126. The PEM 126 of FIG. 3 is shown toinclude a plurality of switches 302 (e.g., IGBTs) configured tocollectively operate as an inverter with first, second, and third phaselegs 316, 318, 320. While the inverter is shown as a three-phaseconverter, the inverter may include additional phase legs. For example,the inverter may be a four-phase converter, a five-phase converter, asix-phase converter, etc. In addition, the PEM 126 may include multipleconverters with each inverter in the PEM 126 including three or morephase legs. For example, the system 300 may control two or moreinverters in the PEM 126. The PEM 126 may further include a DC to DCconverter having high power switches (e.g., IGBTs) to convert a powerelectronics module input voltage to a power electronics module outputvoltage via boost, buck or a combination thereof.

As shown in FIG. 3, the inverter may be a DC-to-AC converter. Inoperation, the DC-to-AC converter receives DC power from a DC power link306 through a DC bus 304 and converts the DC power to AC power. The ACpower is transmitted via the phase currents ia, ib, and ic to drive anAC machine also referred to as an electric machine 114, such as athree-phase permanent-magnet synchronous motor (PMSM) as depicted inFIG. 3. In such an example, the DC power link 306 may include a DCstorage battery to provide DC power to the DC bus 304. In anotherexample, the inverter may operate as an AC-to-DC converter that convertsAC power from the AC machine 114 (e.g., generator) to DC power, whichthe DC bus 304 can provide to the DC power link 306. Furthermore, thesystem 300 may control the PEM 126 in other power electronic topologies.

With continuing reference to FIG. 3, each of the phase legs 316, 318,320 in the inverter includes power switches 302, which may beimplemented by various types of controllable switches. In oneembodiment, each power switch 302 may include a diode and a transistor,(e.g., an IGBT). The diodes of FIG. 3 are labeled D_(a1), D_(a2),D_(b1), D_(b2), D_(c1), and D_(c2) while the IGBTs of FIG. 3 arerespectively labeled S_(a1), S_(a2), S_(b1), S_(b2), S_(c1), and S_(c2).The power switches S_(a1), S_(a2), D_(a1), and D_(a2) are part of phaseleg A of the three-phase converter, which is labeled as the first phaseleg A 316 in FIG. 3. Similarly, the power switches S_(b1), S_(b2),D_(b1), and D_(b2) are part of phase leg B 318 and the power switchesS_(c1), S_(c2), D_(c1), and D_(c2) are part of phase leg C 320 of thethree-phase converter. The inverter may include any number of the powerswitches 302 or circuit elements depending on the particularconfiguration of the inverter. The diodes (D_(xx)) are connected inparallel with the IGBTs (S_(xx)) however, as the polarities are reversedfor proper operation, this configuration is often referred to as beingconnected anti-parallel. A diode in this anti-parallel configuration isalso called a freewheeling diode.

As illustrated in FIG. 3, current sensors CS_(a), CS_(b), and CS_(c) areprovided to sense current flow in the respective phase legs 316, 318,320. FIG. 3 shows the current sensors CS_(a), CS_(b), and CS_(c)separate from the PEM 126. However, current sensors CS_(a), CS_(b), andCS_(c) may be integrated as part of the PEM 126 depending on itsconfiguration. Current sensors CS_(a), CS_(b), and CS_(c) of FIG. 3 areinstalled in series with each of phase legs A, B and C (i.e., phase legs316, 318, 320 in FIG. 3) and provide the respective feedback signalsi_(as), i_(bs), and i_(cs) (also illustrated in FIG. 3) for the system300. The feedback signals i_(as), i_(bs), and i_(cs) may be raw currentsignals processed by logic device (LD) 310 or may be embedded or encodedwith data or information about the current flow through the respectivephase legs 316, 318, 320. Also, the power switches 302 (e.g., IGBTs) mayinclude current sensing capability. The current sensing capability mayinclude being configured with a current mirror output, which may providedata/signals representative of i_(as), i_(bs), and i_(cs). Thedata/signals may indicate a direction of current flow, a magnitude ofcurrent flow, or both the direction and magnitude of current flowthrough the respective phase legs A, B, and C.

Referring again to FIG. 3, the system 300 includes a logic device (LD)or controller 310. The controller or LD 310 can be implemented byvarious types or combinations of electronic devices and/ormicroprocessor-based computers or controllers. To implement a method ofcontrolling the PEM 126, the controller 310 may execute a computerprogram or algorithm embedded or encoded with the method and stored involatile and/or persistent memory 312. Alternatively, logic may beencoded in discrete logic, a microprocessor, a microcontroller, or alogic or gate array stored on one or more integrated circuit chips. Asshown in the embodiment of FIG. 3, the controller 310 receives andprocesses the feedback signals i_(as), i_(bs), and i_(cs) to control thephase currents i_(a), i_(b), and i_(c) such that the phase currentsi_(a), i_(b), and i_(c) flow through the phase legs 316, 318, 320 andinto the respective windings of the electric machine 114 according tovarious current or voltage patterns. For example, current patterns caninclude patterns of phase currents i_(a), i_(b), and i_(c) flowing intoand away from the DC-bus 304 or a DC-bus capacitor 308. The DC-buscapacitor 308 of FIG. 3 is shown separate from the PEM 126. However, theDC-bus capacitor 308 may be integrated as part of the PEM 126.

As shown in FIG. 3, a storage medium 312 (hereinafter “memory”), such ascomputer-readable memory may store the computer program or algorithmembedded or encoded with the method. In addition, the memory 312 maystore data or information about the various operating conditions orcomponents in the PEM 126. For example, the memory 312 may store data orinformation about current flow through the respective phase legs 316,318, 320. The memory 312 can be part of the controller 310 as shown inFIG. 3. However, the memory 312 may be positioned in any suitablelocation accessible by the controller 310.

As illustrated in FIG. 3, the controller 310 transmits at least onecontrol signal 236 to the power converter system 126. The powerconverter system 126 receives the control signal 322 to control theswitching configuration of the inverter and therefore the current flowthrough the respective phase legs 316, 318, and 320. The switchingconfiguration is a set of switching states of the power switches 302 inthe inverter. In general, the switching configuration of the inverterdetermines how the inverter converts power between the DC power link 306and the electric machine 114.

To control the switching configuration of the inverter, the inverterchanges the switching state of each power switch 302 in the inverter toeither an ON state or an OFF state based on the control signal 322. Inthe illustrated embodiment, to switch the power switch 302 to either ONor OFF states, the controller/LD 310 provides the gate voltage (Vg) toeach power switch 302 and therefore drives the switching state of eachpower switch 302. Gate voltages Vg_(a1), Vg_(a2), Vg_(b1), Vg_(b2),Vg_(c1), and Vg_(c2) (shown in FIG. 3) control the switching state andcharacteristics of the respective power switches 302. While the inverteris shown as a voltage-driven device in FIG. 3, the inverter may be acurrent-driven device or controlled by other strategies that switch thepower switch 302 between ON and OFF states. The controller 310 maychange the gate drive for each IGBT based on the rotational speed of theelectric machine 114, the mirror current, or a temperature of the IGBTswitch. The change in gate drive may be selected from a plurality ofgate drive currents in which the change gate drive current isproportional to a change in IGBT switching speed.

As also shown in FIG. 3, each phase leg 316, 318, and 320 includes twoswitches 302. However, only one switch in each of the legs 316, 318, 320can be in the ON state without shorting the DC power link 306. Thus, ineach phase leg, the switching state of the lower switch is typicallyopposite the switching state of the corresponding upper switch.Consequently, a HIGH state of a phase leg refers to the upper switch inthe leg in the ON state with the lower switch in the OFF state.Likewise, a LOW state of the phase leg refers to the upper switch in theleg in the OFF state with the lower switch in the ON state. As a result,IGBTs with current mirror capability may be on all IGBTs, a subset ofIGBTs (e.g., S_(a1), S_(b1), S_(c1)) or a single IGBT.

Two situations can occur during an active state of the three-phaseconverter example illustrated in FIG. 2: (1) two phase legs are in theHIGH state while the third phase leg is in the LOW state, or (2) onephase leg is in the HIGH state while the other two phase legs are in theLOW state. Thus, one phase leg in the three-phase converter, which maybe defined as the “reference” phase for a specific active state of theinverter, is in a state opposite to the other two phase legs, or“non-reference” phases, that have the same state. Consequently, thenon-reference phases are either both in the HIGH state or both in theLOW state during an active state of the inverter.

FIG. 4 is an example graphical representation of a profile 400 of a gatecurrent 402 with respect to time 404. Here, the profile 400 includes anincrease of the gate current 402 at time 406 to a first gate currentlevel 408 (I_(g1)). The gate current 402 is substantially maintaineduntil a transition time 410 at which the first gate current level 408 isreduced to a second gate current level 412 (I_(g2)). The time the gatecurrent is maintained at the first gate current level 408 (I_(g1)) isreferred to as a high time or initial gate current drive time (t1). Thefirst gate current level 408 may be based on many factors includingdevice characteristics, operating conditions, and load characteristics.The first gate current level 408 may be explained by the trade-offbetween performance and cost. Generally, the higher the first gatecurrent level 408 is, the lower the loss is. However, the gate currentis limited by gate driver capabilities, as the cost associated with thegate driver is typically directly proportional with the gate drivercapabilities. The device characteristics include a gate resistance, agate capacitance, a threshold voltage, a max collector current, a diodeforward current, and other IGBT characteristics. The operatingconditions include a temperature, a switching speed, a PWM duty cycle, asupply voltage, and a vehicle speed. The load characteristics include aninductance, a resistance, a max current, rotational speed, potentialenergy, kinetic energy, and other electrical or electro-mechanicalcharacteristics. The second gate current level 412 may be based on manyfactors including the factors used for the first gate current level 408.The second gate current level 412 is determined by diode voltage spikein the worst operating case (i.e., maximum dc bus voltage and maximumload current). The second gate current level 412 (I_(g2)) is maintainedfrom time 410 until time 414. The time the gate current is maintained atthe second gate current level 412 (I_(g2)) is referred to as a low timeor second gate current drive time (t2).

The transition time 410 may be implemented by a variety of methodsincluding a pre-set driving profile, an open-loop control system, aclosed-loop control system, an analog control system, a digital controlsystem, an adaptive control system, a fuzzy logic control system, and aneural network system. In a first embodiment, the transition time may bein response to an integral over a time period, also referred to as atime integral, of a load driven by the IGBT exceeding a referencethreshold. The device characteristics may include a gate resistance, ora gate capacitance. The operating conditions may include a switchingspeed or an IGBT temperature. The load characteristics may include aload inductance, a load capacitance, or a load resistance. In yetanother embodiment, the transition time may be in response to a timeintegral of a voltage measured from the emitter to the collector of theIGBT exceeding a reference threshold.

FIG. 5 is an example flow diagram 500 of a method to drive a gate of anIGBT. In operation 502, a controller receives signal to turn on an IGBTdevice after the controller proceeds to operation 504. In operation 504,the controller determines an initial gate current level. Generally, aturn-on switching loss is inversely proportional to the gate currentlevel. However, the initial gate current may also be limited by gatedriver capabilities, as the cost associated with the gate driver istypically directly proportional with the gate driver capabilities. Forexample, a gate driver with higher driving capabilities is typicallymore expensive than a gate driver with lower driving capabilities.

In operation 504, the controller selects the initial gate current basedon multiple factors including a switching loss of the IGBT device, agate driver loss, and a capability of a gate driving circuit. Once theinitial gate current is selected, the controller proceeds to operation506. For example, the initial gate current level may be I_(g1) from FIG.4.

In operation 506, the controller monitors operating conditions of theIGBT device. The operating conditions monitored include a voltage acrossthe collector and emitter of the IGBT device, a current flowing into acollector of the IGBT device, a voltage across a load coupled witheither a collector or an emitter of the IGBT, a current flowing to theload coupled with either the collector or the emitter of the IGBT or atemperature of the IGBT device. The monitoring may include measuring acharacteristic and calculating a time integral of a component. Forexample, an inverter to drive an electric machine may have an IGBTdevice coupled with each phase of the electric machine. The controllerof the IGBT device may calculate or receive as an input the timeintegral of a voltage across a phase of the electric machine coupledwith the IGBT. Typically, the voltage across IGBT is directly measured.The different between the voltage across IGBT and the voltage across thephase of the electric machine equals the dc bus voltage, along with somevoltage drop on the parasitic inductances.

In operation 508, the controller compares the sensed signal fromoperation 506 with a trigger value. For example, the controller maycompare the time integral of the voltage across the phase of theelectric machine coupled with the IGBT with a predetermined value. Thelength of the initial gate current is important as a longer initial gatecurrent drive time leads to a higher IGBT di/dt and a higher voltagespike across the freewheeling diode. Here, initial gate current drivetime is gradually increased until the time integral of the voltage ofthe load (such as a phase of the electric machine for an inverter or aninductor load for a DC-DC converter) exceeds a predetermined threshold.The predetermined threshold is a point in which a voltage across thefreewheeling diode spikes to a limit less than a breakdown voltage ofthe IGBT. This voltage is dependent upon many conditions such as asupply voltage for the electric machine or a supply voltage for theDC-DC converter. Upon reaching or exceeding the predetermined threshold,the controller will proceed to operation 510. While the time integral isless than the predetermined threshold, the controller will branch backto operation 506.

In operation 510, the controller decrements the gate current to a lowergate current level and monitors the IGBT operation. For example, in FIG.4, the gate current is reduced from I_(g1) to I_(g2), operation 510 mayinclude the transition from I_(g1) to I_(g2), followed by monitoring theIGBT operation until time 414. In practice, the initial gate currentlevel is usually larger than the lower gate current level. The lowergate current level is selected to avoid the excessive voltage overshootacross the freewheeling diode during multiple operating ranges.Typically a worst case operating condition (e.g., maximum dc busvoltage) is selected to determine the lower gate current level. Once abaseline is determined, the lower gate current level may be increasedgradually until a voltage spike across the freewheeling diode exceeds alimit. Therefore, the lower gate current level may be determined basedon a DC bus voltage, or a battery voltage.

In operation 512, the controller compares an amount of charge applied tothe gate of the IGBT, and if the charge applied to the gate equals afully enhanced charge level, the controller may reduce the gate currentto a leakage gate current level and proceed to operation 514. Inoperation 514, the controller stops driving a gate current to the IGBTdevice.

FIG. 6 is an example schematic of a IGBT gate drive circuit 600. Here, agate driver 602 is configured to drive a gate of an IGBT 604. Thecircuit 600 includes a comparator 606 to switch states in response to aninput exceeding a threshold. The input to the comparator may be coupledwith a collector of the IGBT 604 by a resistor 608. In response to avoltage greater than a threshold voltage applied to a gate of the IGBT604, the IGBT 604 begins conducting a current between the emitter andcollector of the IGBT 604. One circuit to measure a time integral of thevoltage between the emitter and collector of the IGBT 604 is to enablethe MOSFET 612 thus allow a current to flow through the resistor 608 tothe capacitor 614 and the resistor 616. When the time integral exceedsthe threshold indicated by Vref, the comparator 606 will output a signalto the gate driver controller 602 to enable the transition from Ig1 toIg2. Based on the desired operation, a gate resistor 610 may be includedto soften the turn on and transition. This exemplary circuit isconfigured to determine the time integral for Vce, however, a similarcircuit may be used to determine the time integral for other componentssuch as the load. An advantage of operating based on the time integralof the load includes adaptation of current operating conditions such asrotational speed, kinetic energy of the electric machine, potentialenergy of the electric machine. Likewise, a similar circuit may be usedin a VVC to determine a time integral for a power inductor (e.g. inputinductor 214). This adaptive system will adjust the point of thetransition time 410, for example, the Ig1 pulse length will increase asthe dc bus voltage decreases and may decrease as the dc bus voltageincreases.

The processes, methods, or algorithms disclosed herein can bedeliverable to/implemented by a processing device, controller, orcomputer, which can include any existing programmable electronic controlunit or dedicated electronic control unit. Similarly, the processes,methods, or algorithms can be stored as data and instructions executableby a controller or computer in many forms including, but not limited to,information permanently stored on non-writable storage media such asRead Only Memory (ROM) devices and information alterably stored onwriteable storage media such as floppy disks, magnetic tapes, CompactDiscs (CDs), Random Access Memory (RAM) devices, and other magnetic andoptical media. The processes, methods, or algorithms can also beimplemented in a software executable object. Alternatively, theprocesses, methods, or algorithms can be embodied in whole or in partusing suitable hardware components, such as Application SpecificIntegrated Circuits (ASICs), Field-Programmable Gate Arrays (FPGAs),state machines, controllers or other hardware components or devices, ora combination of hardware, software and firmware components.

While exemplary embodiments are described above, it is not intended thatthese embodiments describe all possible forms encompassed by the claims.The words used in the specification are words of description rather thanlimitation, and it is understood that various changes can be madewithout departing from the spirit and scope of the disclosure. Aspreviously described, the features of various embodiments can becombined to form further embodiments of the invention that may not beexplicitly described or illustrated. While various embodiments couldhave been described as providing advantages or being preferred overother embodiments or prior art implementations with respect to one ormore desired characteristics, those of ordinary skill in the artrecognize that one or more features or characteristics can becompromised to achieve desired overall system attributes, which dependon the specific application and implementation. These attributes mayinclude, but are not limited to cost, strength, durability, life cyclecost, marketability, appearance, packaging, size, serviceability,weight, manufacturability, ease of assembly, etc. As such, embodimentsdescribed as less desirable than other embodiments or prior artimplementations with respect to one or more characteristics are notoutside the scope of the disclosure and can be desirable for particularapplications.

What is claimed is:
 1. A vehicle comprising: an electric machine; anIGBT having a gate, an emitter, and a collector, configured to flow anelectric charge through a phase of the electric machine; and a gatedriver configured to flow current onto the gate at a first level, and inresponse to a time integral of a voltage across the phase exceeding apredetermined level, transition from the first level to a second levelless than the first level.
 2. The vehicle of claim 1, wherein thepredetermined level is based on an electric potential of the electriccharge.
 3. The vehicle of claim 1, wherein the first level is based on agate capacitance of the IGBT, and an electric potential of the electriccharge.
 4. The vehicle of claim 1 further comprising a freewheelingdiode coupled anti-parallel with the IGBT, wherein the predeterminedlevel is derived from an electric potential of the electric charge suchthat an overshoot voltage across the freewheeling diode does not exceeda diode limit.
 5. The vehicle of claim 4, wherein the second level isbased on a rate of change of collector current of the IGBT such that amaximum diode voltage does not exceed a breakdown threshold.
 6. Thevehicle of claim 1, wherein the time integral is for a period beginningwhen the gate driver receives an IGBT turn-on signal and ending when thegate driver turns off the IGBT.
 7. The vehicle of claim 1, wherein thefirst level is based on a parasitic inductance associated with the gateof the IGBT, and a temperature of the IGBT.
 8. A method of controllingan IGBT of a power system comprising: by a gate driver, flowing acurrent at a first level onto a gate of an IGBT; and in response to atime integral of a voltage across the IGBT exceeding a predeterminedthreshold, transitioning the current from the first level to a secondlevel less than the first level.
 9. The method of claim 8, wherein thefirst level is based on a gate capacitance of the IGBT.
 10. The methodof claim 8, wherein the second level is based on a rate of change ofcollector current of the IGBT such that a reverse bias diode voltageacross a freewheeling diode, coupled anti-parallel with the IGBT, doesnot exceed a breakdown threshold.
 11. The method of claim 8, wherein thetime integral is for a period beginning when the gate driver receives anIGBT turn-on signal and ending when the gate driver turns off the IGBT.12. The method of claim 8, wherein the predetermined threshold isderived from an electric potential associated with the current such thatan overshoot voltage across a freewheeling diode coupled anti-parallelwith the IGBT does not exceed a diode limit.
 13. A vehicle powertraincomprising: an IGBT having a gate, an emitter and a collector; and agate driver configured to flow current onto the gate at a first level,and in response to a time integral of resulting collector to emittervoltage exceeding a predetermined level, transition from the first levelto a second level less than the first level.
 14. The vehicle powertrainof claim 13, wherein the first level is based on a gate capacitance ofthe IGBT and a parasitic inductance of the IGBT.
 15. The vehiclepowertrain of claim 13 further comprising a freewheeling diode coupledanti-parallel with the IGBT, wherein the predetermined level is derivedfrom a voltage associated with the current such that an overshootvoltage across the freewheeling diode does not exceed a diode limit. 16.The vehicle powertrain of claim 15, wherein the second level is based ona rate of change of collector current of the IGBT such that a maximumdiode voltage does not exceed a breakdown threshold.
 17. The vehiclepowertrain of claim 13, wherein the time integral is for a periodbeginning when the gate driver receives an IGBT turn-on signal andending when the gate driver turns off the IGBT.
 18. The vehiclepowertrain of claim 13, wherein the first level is based on a parasiticinductance associated with the gate of the IGBT, and a temperature ofthe IGBT.